Since the adoption of the ATSC digital television (DTV) standard in 1996, there has been an ongoing effort to improve the design of receivers built for the ATSC DTV signal. The primary obstacle that faces designers in designing receivers so that they achieve good reception is the presence of multipath interference in the channel. Such multipath interference affects the ability of the receiver to recover signal components such as the carrier and symbol clock. Therefore, designers add equalizers to receivers in order to cancel the effects of multipath interference and thereby improve signal reception.
The broadcast television channel is a relatively severe multipath environment due to a variety of conditions that are encountered in the channel and at the receiver. Strong interfering signals may arrive at the receiver both before and after the largest amplitude signal. In addition, the signal transmitted through the channel is subject to time varying channel conditions due to the movement of the transmitter and signal reflectors, airplane flutter, and, for indoor reception, people walking around the room. If mobile reception is desired, movement of the receiver must also be considered.
The ATSC DTV signal uses a 12-phase trellis coded 8-level vestigial sideband (usually referred to as 8T-VSB or, more simply, as 8-VSB) as the modulation method. There are several characteristics of the 8-VSB signal that make it special compared to most linear modulation methods (such as QPSK or QAM) that are currently used for wireless transmission. For example, 8-VSB data symbols are real and have a signal pulse shape that is complex. Only the real part of the complex pulse shape is a Nyquist pulse. Therefore, the imaginary part of the complex pulse shape contributes intersymbol interference (ISI) when the channel gain seen by the equalizer is not real, even if there is no multipath.
Also, due to the low excess bandwidth, the signal is nearly single sideband. As a result, symbol rate sampling of the complex received signal is well above the Nyquist rate. Symbol rate sampling of the real or imaginary part of the received signal is just below the Nyquist rate.
Because the channel is not known a priori at the receiver, the equalizer must be able to modify its response to match the channel conditions that it encounters and to adapt to changes in those channel conditions. To aid in the convergence of an adaptive equalizer to the channel conditions, the field sync segment of the frame as defined in the ATSC standard may be used as a training sequence for the equalizer. But when equalization is done in the time domain, long equalizers (those having many taps) are required due to the long channel impulse responses that characterize the channel. Indeed, channels are often characterized by impulse responses that can be several hundreds of symbols long.
The original Grand Alliance receiver used an adaptive decision feedback equalizer (DFE) with 256 taps. The adaptive decision feedback equalizer was adapted to the channel using a standard least mean square (LMS) algorithm, and was trained with the field sync segment of the transmitted frame. Because the field sync segment is transmitted relatively infrequently (about every 260,000 symbols), the total convergence time of this equalizer is quite long if the equalizer only adapts on training symbols prior to convergence.
In order to adapt equalizers to follow channel variations that occur between training sequences, it had been thought that blind and decision directed methods could be added to equalizers. However, when implemented in a realistic system, these methods may require several data fields to achieve convergence, and convergence may not be achieved at all under difficult multipath conditions.
In any event, because multipath signals in the broadcast channel may arrive many symbols after the main signal, the decision feedback equalizer is invariably used in 8-VSB applications. However, it is well known that error propagation is one of the primary drawbacks of the decision feedback equalizer. Therefore, under severe multipath conditions, steps must be taken to control the effect of error propagation.
In a coded system, it is known to insert a decoder into the feedback path of the decision feedback equalizer to use the tentative decision of the decoder in adapting the equalizer to channel conditions. This method, or a variant of it, is applicable to the 8-VSB signal by way of the output of the trellis decoder. As discussed above, the ATSC DTV signal is a 12-phase trellis coded digital vestigial sideband signal with 8 signal levels known as 8T-VSB.
In ATSC DTV systems, data is transmitted in frames as shown in FIG. 1. Each frame contains two data fields, each data field contains 313 segments, and each segment contains 832 symbols. The first four of these symbols in each segment are segment sync symbols having the sequence [+5, −5, −5, +5].
The first segment in each field is a field sync segment. As shown in FIG. 2, the field sync segment comprises the four segment sync symbols discussed above followed by a pseudo-noise sequence having a length of 511 symbols (PN511) followed in turn by three pseudo-noise sequences each having a length of 63 symbols (PN63). Like the segment sync symbols, all four of the pseudo-noise sequences are composed of symbols from the set {+5, −5}. In alternate fields, the three PN63 sequences are identical; in the remaining fields, the center PN63 sequence is inverted. The pseudo-noise sequences are followed by 128 symbols, which are composed of various mode, reserved, and precode symbols.
Because the first 704 symbols of each field sync segment are known, these symbols, as discussed above, may be used as a training sequence for an adaptive equalizer. All of the three PN63 sequences can be used only when the particular field being transmitted is detected so that the polarity of the center sequence is known. The remaining data in the other 312 segments comprises trellis coded 8-VSB symbols. This data, of course, is not known a-priori by the receiver.
A transmitter 10 for transmitting the 8T-VSB signal is shown at a very high level in FIG. 3. The transmitted baseband 8T-VSB signal is generated from interleaved Reed-Solomon coded data. After trellis coding by a trellis encoder 12, a multiplexer 14 adds the segment sync symbols and the field sync segment to the trellis coded data at the appropriate times in the frame. A pilot inserter 16 then inserts a pilot carrier by adding a DC level to the baseband signal, and a modulator 18 modulates the resulting symbols. The modulated symbols are transmitted as a vestigial sideband (VSB) signal at a symbol rate of 10.76 MHz.
FIG. 4 shows the portions of the transmitter and receiver relevant to the analysis presented herein. The transmitted signal has a raised cosine spectrum with a nominal bandwidth of 5.38 MHz and an excess bandwidth of 11.5% of the channel centered at one-fourth of the symbol rate (i.e., 2.69 MHz). Thus, the transmitted pulse shape q(t) (block 20, FIG. 4) is complex and is given by the following equation:q(t)=ejπFst/2qRRC(t)  (1)where Fs is the symbol frequency, and qRRC(t) is a real square root raised cosine pulse with an excess bandwidth of 11.5% of the channel. Thus, the pulse q(t) is a complex root raised cosine pulse.
The baseband transmitted signal waveform of data rate 1/T symbols/sec is represented by the following equation:
                              s          ⁡                      (            t            )                          =                              ∑            k                    ⁢                                    I              k                        ⁢                          q              ⁡                              (                                  t                  -                  kT                                )                                                                        (        2        )            where {Ik∈A≡{α1, . . . α8}⊂R1} is the transmitted data sequence, which is a discrete 8-ary sequence taking values on the real 8-ary alphabet A. The function q(t) is the transmitter's pulse shaping filter of finite support [−Tq/2, Tq/2]. The overall complex pulse shape at the output of the matching filter in the receiver is denoted p(t) and is given by the following equation:p(t)=q(t)*q*(−t)  (3)where q*(−t) (block 22, FIG. 4) is the receiver matched filter impulse response.
Although it is not required, it may be assumed for the sake of simplifying the notation that the span Tq of the transmit filter and the receive filter is an integer multiple of the symbol period T; that is, Tq=NqT=2LqT, and Lq is a real integer greater than zero. For the 8-VSB system, the transmitter pulse shape is the Hermitian symmetric root raised cosine pulse, which implies that q(t)=q*(−t). Therefore, q[n]≡q(t)|t=nT is used below to denote both the discrete transmit filter and discrete receive filter.
The physical channel between the transmitter and the receiver is denoted c(t) (block 24, FIG. 4). The concatenation of p(t) and the channel is denoted h(t) and is given by the following equation:h(t,τ)=q(t)*c(t,τ)*q*(−t)=p(t)*c(t,τ)  (4)The physical channel c(t,τ) is generally described as a time varying channel by the following impulse response:
                              c          ⁡                      (                          t              ,              τ                        )                          =                              ∑                          k              =                              -                                  L                  ha                                                                    L              hc                                ⁢                                                    c                k                            ⁡                              (                τ                )                                      ⁢                          δ              ⁡                              (                                  t                  -                                      τ                                          k                      ⁢                                                                                                                                          )                                                                        (        5        )            where {ck(τ)}⊂C1, where −Lha≦k≦Lhc, t, τ∈R, and {τk} denote the multipath delays, or the time of arrivals (TOA), and where δ(t) is the Dirac delta function. It is assumed that the time variations of the channel are slow enough that c(t,τ)=c(t). Thus, the channel is assumed to be a fixed (static) inter-symbol interference channel throughout the training period such that ck(τ)=ck, which in turn implies the following equation:
                              c          ⁡                      (            t            )                          =                              ∑                          k              =                              -                                  L                  ha                                                                    L              hc                                ⁢                                    c              k                        ⁢                          δ              ⁡                              (                                  t                  -                                      τ                    k                                                  )                                                                        (        6        )            for 0≦t≦LnT, where Ln is the number of training symbols, and the summation indices Lha and Lhc refer to the number of maximum anti-causal and causal multipath delays, respectively.
In general, ck={tilde over (c)}ke−j2πfcτk where {tilde over (c)}k is the amplitude of the k′th multipath, and fc is the carrier frequency. It is also inherently assumed that τk<0 for −Lha≦k≦−1, τ0=0, and τk>0 for 1≦k≦Lhc. The multipath delays τk are not assumed to be at integer multiples of the sampling period T.
Equations (4) and (6) may be combined according to the following equation (where the τ index has been dropped):
                              h          ⁡                      (            t            )                          =                                            p              ⁡                              (                t                )                                      *                          c              ⁡                              (                t                )                                              =                                    ∑                              -                                  L                  ha                                                            L                hc                                      ⁢                                          c                k                            ⁢                              p                ⁡                                  (                                      t                    -                                          τ                      k                                                        )                                                                                        (        7        )            
Because both p(t) and c(t) are complex valued functions, the overall channel impulse response h(t) is also complex valued. By using the notation introduced herein, the matched filter output y(t) in the receiver is given by the following equation:
                              y          ⁡                      (            t            )                          =                                            (                                                ∑                  k                                ⁢                                  δ                  ⁡                                      (                                          t                      -                      kT                                        )                                                              )                        *                          h              ⁡                              (                t                )                                              +                      v            ⁡                          (              t              )                                                          (        8        )            wherev(t)=η(t)*q*(−t)  (9)denotes the complex (colored) noise process after the pulse matched filter (denoted by block 25, FIG. 4), with η(t) being a zero-mean white Gaussian noise process with spectral density ση2 per real and imaginary part. The matched filter output y(t) can also be written in terms of its real and imaginary parts as y(t)=yI(t)+jyQ(t).
Sampling the matched filter output y(t) (sampler 26, FIG. 4) at the symbol rate produces the discrete time representation of the overall communication system according to the following equation:
                                                        y              ⁡                              [                n                ]                                      ≡                          y              ⁡                              (                t                )                                              ⁢                      |                          t              =              nT                                      =                                            ∑              k                        ⁢                                          I                k                            ⁢                              h                ⁡                                  [                                      n                    -                    k                                    ]                                                              +                      v            ⁡                          [              n              ]                                                          (        10        )            Prior art equalizers have known problems previously discussed, such as having difficulty in converging under severe multipath conditions.
The present invention provides a novel technique to provide improved convergence time of equalizers and/or to solve other problems associated with equalizers.